Jitter noise detector

ABSTRACT

A noise detection circuit includes a first transistor configured to receive a delayed version of a clock signal; a second transistor configured to receive a delayed version of a reference clock signal; and a latch circuit, coupled to the first transistor at a first node and coupled to the second transistor at a second node, and configured to latch logic states of voltage levels at the first and second nodes, respectively, based on whether a timing difference between transition edges of the clock signal and the reference clock signal exceeds a pre-defined timing offset threshold.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation application of U.S. patentapplication Ser. No. 15/944,217, filed Apr. 3, 2018, which claimspriority to U.S. Provisional Patent Application No. 62/525,656, filed onJun. 27, 2017, each of which are incorporated by reference herein intheir entireties.

BACKGROUND

In electronic and/or telecommunication applications, jitter is a timedeviation from a true periodicity of a presumably periodic signal. Amongvarious causes of the jitter are electromagnetic interference (EMI) andcrosstalk with other periodic or non-periodic signals. Such jitter istypically considered as a noise effect in a circuit, device or system.The jitter generally cause various issues for a respective circuit,device or system such as, for example, causing a display monitor toflicker, disadvantageously affecting an ability of a processor of adesktop or server to perform as originally intended operation, inducingclicks or other undesired effects in audio signals, loss of transmitteddata between network devices, etc. Thus, there exists a need for atechnique to accurately and quickly detect the presence of jitter in acircuit, device or system.

BRIEF DESCRIPTION OF THE DRAWINGS

Aspects of the present disclosure are best understood from the followingdetailed description when read with the accompanying figures. It isnoted that various features are not necessarily drawn to scale. In fact,the dimensions of the various features may be arbitrarily increased orreduced for clarity of discussion.

FIG. 1 illustrates an exemplary circuit diagram of a p-type jitterdetection (pJD) circuit, in accordance with some embodiments.

FIG. 2A illustrates an exemplary circuit diagram of a tuning circuit ofthe pJD circuit of FIG. 1, in accordance with some embodiments.

FIG. 2B illustrates another exemplary circuit diagram of the tuningcircuit of the pJD circuit of FIG. 1, in accordance with someembodiments.

FIG. 3 illustrates exemplary waveforms of plural signals to operate thepJD circuit of FIG. 1, in accordance with some embodiments.

FIG. 4 illustrates an exemplary flow chart of a method to operate thepJD circuit of FIG. 1, in accordance with some embodiments.

FIG. 5 illustrates a exemplary circuit diagram of an n-type jitterdetection (nJD) circuit, in accordance with some embodiments.

FIG. 6 illustrates exemplary waveforms of plural signals to operate thenJD circuit of FIG. 5, in accordance with some embodiments.

FIG. 7 illustrates an exemplary flow chart of a method to operate thenJD circuit of FIG. 5, in accordance with some embodiments.

DETAILED DESCRIPTION OF EXEMPLARY EMBODIMENTS

The following disclosure describes various exemplary embodiments forimplementing different features of the subject matter. Specific examplesof components and arrangements are described below to simplify thepresent disclosure. These are, of course, merely examples and are notintended to be limiting. For example, it will be understood that when anelement is referred to as being “connected to” or “coupled to” anotherelement, it may be directly connected to or coupled to the otherelement, or one or more intervening elements may be present.

The present disclosure provides various embodiments of a jitterdetection circuit that can accurately detect a presence of jitter in aclock signal. More particularly, in some embodiments, the disclosedjitter detection circuit compares respective transition edges (e.g.,rising edges, falling edges, etc.) of the clock signal and a referenceclock signal by using either a p-type or an n-type jitter detectioncircuit so as to determine whether the jitter is present in the clocksignal in a real-time fashion. Moreover, in some embodiments, the p-typeand n-type jitter detection circuits each includes a tuning circuit thatallows the respective p-type and n-type jitter detection circuits totune respective jitter detection sensitivities. In some embodiments,such a jitter detection sensitivity may be referred to as a minimumquantified amount of the jitter that can be detected, for example, aminimum timing offset window of either the rising edges or the fallingedges between the clock signal and reference clock signal.

FIG. 1 illustrates an exemplary circuit diagram of a p-type jitterdetection circuit (hereinafter “pJD circuit”) 100, in accordance withsome embodiments. As mentioned above, the pJD circuit 100 is configuredto compare respective transition edges between a clock signal (e.g.,101) and a reference clock signal (e.g., 103) so as to determine whetherthe clock signal 101 contains jitter that exceeds a pre-definedthreshold (e.g., a pre-defined timing offset window T_(os), which willbe discussed in further detail below with respect to FIG. 3). If so, thepJD circuit 100 may output signal 105 at a high logic state (hereinafter“HIGH”). On the other hand, if no jitter is detected or the jitter inthe clock signal 101 does not exceed the pre-defined threshold, the pJDcircuit 100 may output the signal 105 at a low logic state (hereinafter“LOW”).

In some embodiments, the pJD circuit 100 is configured to comparerespective “rising” edges of the clock signal 101 and the referenceclock signal 103. The clock signal 101 may be generated by a clockgeneration circuit, for example, a phase-locked-loop (PLL) circuitintegrated in a bigger system circuit (e.g., a system-on-chip (SoC)circuit, an application-specific integrated circuit (ASIC), etc.). Thereference clock signal 103 may be provided by an external crystalcircuit, which is generally considered as a relatively reliable clockgeneration source, thus making the reference clock signal 103 a reliablereference. In some other embodiments, the reference clock 103 can beprovided by either delaying the clock signal 101 by a pre-defined periodof time or from another low-noise PLL, even off-chip instruments. Theclock generation circuit, which provides the clock signal 101, may beconfigured to provide one or more synchronous or asynchronousfunctionalities to the bigger system circuit. Thus, by coupling thedisclosed pJD circuit 100 to such a bigger system circuit, the clocksignal 101 may be accurately monitored in a real-time manner, which willbe described in further detail below with respect to FIG. 3.

Referring still to FIG. 1, in some embodiments, the pJD circuit 100includes a first delay circuit 110, a second delay circuit 112, a logicgate 114, transistors 116, 118, 120, 122, 124, 126, 128, and 130,inverters 132 and 134, a logic gate 136, and a tuning circuit 138. Insome embodiments, the first and second delay circuits 110 and 112 mayeach include a plurality of serially coupled buffers, inverters, or thelike (not shown). The first delay circuit 110 is configured to receivethe clock signal 101 and provide a delayed version of the clock signal,e.g., 101′, and the second delay circuit 112 is configured to receivethe reference clock signal 103 and provide a delayed version of theclock signal, e.g., 103′. In some embodiments, the logic gate 114 of thepJD circuit 100 may include a NAND logic gate that is configured toperform a NAND logic function on the clock signal 101 and the referenceclock signal 103 so as to provide a control signal 114′ based on aNAND'ed result of logic states of the clock signal 101 and the referenceclock signal 103.

In some embodiments, the transistors 116, 124, 126, 128, and 130 may beeach implemented by an n-type metal-oxide-semiconductor (NMOS)field-effect-transistor (FET), and the transistors 118, 120, and 122 maybe each implemented by a p-type metal-oxide-semiconductor (PMOS)field-effect-transistor (FET). However, it is noted that the transistors116 to 130 may each be implemented as any of various types oftransistors (e.g., a bipolar junction transistor (BJT), a high-electronmobility transistor (HEMT), etc.) while remaining within the scope ofthe present disclosure.

More specifically, the transistors 116 and 118 are commonly coupled to afirst supply voltage 107 (e.g., Vdd) at respective drain and source, andgated by the control signal 114′. The transistor 120 is coupled to thetransistor 116's source by its respective source, and gated by thedelayed clock signal 101′. Similarly, the transistor 122 is coupled tothe transistor 118's drain by its respective source, and gated by thedelayed reference clock signal 103′. And the transistor 118's drain iscoupled to the transistor 116's source. The transistors 124 and 126 arecoupled to a drain of the transistor 120 by their respective drains at acommon node “X,” and to a second supply voltage 109 (e.g., Vss orground) by their respective sources. In some embodiments, the transistor124 is gated by the control signal 114′. Similarly, the transistors 128and 130 are coupled to a drain of the transistor 122 by their respectivedrains at a common node “Y,” and to the second supply voltage 109 (e.g.,Vss or ground) by their respective sources. In some embodiments, thetransistor 130 is gated by the control signal 114′.

More specifically, in some embodiments, the transistors 126 and 128 arecross-coupled to each other. That is, a gate of the transistor 126 iscoupled to the drain of the transistor 128 and a gate of the transistor128 is coupled to the drain of the transistor 126 so as to allow thetransistors 126 and 128 to function as a latch circuit, which will bediscussed in further detail below with respect to FIG. 3.

In some embodiments, the inverters 132 and 134 are configured to receivesignals present at nodes X and Y (hereinafter “signal 131” and “signal133”), respectively, as respective input signals, and provide respectivelogically inverted signals 135 and 137. The signals 135 and 137 arereceived by the logic gate 136, which may be implemented as an XOR logicgate in some embodiments. The logic gate 136 is configured to perform anXOR logic function on the signals 135 and 137 so as to provide thesignal 105 whose logic state is determined based on an XOR'ed result oflogic states of the signals 135 and 137.

An exemplary circuit diagram of the tuning circuit 138 is illustrated inFIG. 2A. In some embodiments, the tuning circuit 138 includes one ormore capacitors 202, 204, and 206 coupled between the nodes X and Y byrespective switches 208, 210, and 212. More specifically, the capacitor202 includes two conductive plates 202-1 and 202-2, wherein oneconductive plate (e.g., 202-1) is coupled to the node Y and the otherconductive plate (e.g., 202-2) is coupled to the node X through theswitch 208; the capacitor 204 includes two conductive plates 204-1 and204-2, wherein one conductive plate (e.g., 204-1) is coupled to the nodeY and the other conductive plate (e.g., 204-2) is coupled to the node Xthrough the switch 208; and the capacitor 206 includes two conductiveplates 206-1 and 206-2, wherein one conductive plate (e.g., 206-1) iscoupled to the node Y and the other conductive plate (e.g., 206-2) iscoupled to the node X through the switch 208.

According to some embodiments, each of the switches 208, 210 and 212 maybe selectively turned on/off to tune the jitter detection sensitivity,i.e., the pre-defined timing offset window T_(os), of the pJD circuit100. More specifically, when more switches are turned on, morecapacitors are electrically coupled between the nodes X and Y, whichcauses the timing offset window T_(os) to become wider. Conversely, whenless switches are turned on, less capacitors are electrically coupledbetween the nodes X and Y, which causes the timing offset window T_(os)to become narrower. As will be discussed in further detail below, such atiming offset window T_(os) may be used to determine whether the logicstates of the signals 131 (i.e., a voltage level at the node X) and 133(i.e., a voltage level at the node Y) can be “latched” by the coupledlatch circuit formed by the transistors 126 and 128. Although only threecapacitors 202, 204 and 206 (and corresponding switches 208, 210 and210) are shown in the illustrated embodiment of FIG. 2, it is understoodthat any desired number of capacitors (and corresponding switches) maybe included in the tuning circuit 138.

Another exemplary circuit diagram of the tuning circuit 138 isillustrated in FIG. 2B, which is herein referred to as tuning circuit138′, for purposes of clarity of illustration. In some embodiments, thetuning circuit 138′ is substantially similar to the circuit diagram ofthe tuning circuit 138 shown in FIG. 2A except that the tuning circuit138′ further includes capacitors 208′, 210′ and 212′. In someembodiments, the capacitor 208′ is coupled between the conductive plate202-1 and the node Y; the capacitor 210′ is coupled between theconductive plate 204-1 and the node Y; and the capacitor 212′ is coupledbetween the conductive plate 206-1 and the node Y. Each of thecapacitors 208′, 210′, and 212′ are substantially similar to thecapacitors 208, 210, and 212, respectively, in terms of functionalityand configuration such that discussions of the capacitors 208′, 210′,and 212′ are not repeated here.

FIG. 3 illustrates exemplary waveforms of signals 101, 103, 114′, 101′,103′, 131, 133, and 105 to operate the pJD circuit 100 of FIG. 1, inaccordance with some embodiments. Each waveform of the signals 101, 103,114′, 101′, 103′, 131, 133, and 105 illustrated in FIG. 3 varies betweenHIGH and LOW over time.

As mentioned above, jitter is a deviation from a true periodicity of apresumably periodic signal. In some embodiments, the reference clocksignal 103 may be used as the “presumably periodic signal” that is usedto examine the clock signal 101 and to determine whether a deviation ofthe clock signal 101 from the presumably periodic signal 103 exceeds thepre-defined timing offset window T_(os). In some embodiments, when theclock signal 101 contains jitter (i.e., has a deviation) that exceedsthe pre-defined timing offset window (i.e., an intolerable amount ofjitter) on its respective rising edge, the pJD circuit 100 may pull thesignal 105 to HIGH, as mentioned above. FIG. 3 illustrates a scenariowhere the clock waveform signal 101 contains jitter that exceeds apredetermined threshold, which is detected by the pJD circuit 100, andthe corresponding signals that are used or generated by the pJD circuit100 (i.e., signals 114′, 101′, 103′, 131, 133 and 105).

As shown in FIG. 3, the clock signal 101's rising edge 101 r deviatesfrom the reference clock signal 103's rising edge 103 r. Morespecifically, the rising edge 101 r occurs “ΔT” ahead of the rising edge103 r. Alternative stated, the rising edges 101 r and 103 r have atiming difference ΔT from each other. As described above, the logic gate114 performs the NAND logic function on the clock signal 101 and thereference clock signal 103. As known in the art, only when both thesignals 101 and 103 transition to HIGH, the logic gate 114 can outputthe control signal 114′ as LOW.

Prior to time “t0,” the control signal 114′ is at HIGH, and at time t0,the control signal 114′ remains at HIGH because the logic states of thesignals 101 and 103 are at LOW. It is noted that the transistors 116,118, 124, and 130 are all gated by the signal 114′. Accordingly, whenthe control signal 114′ is at HIGH, the “NMOS” transistors 116, 124, and130 are turned on, and the “PMOS” transistor 118 is turned off. In someembodiments, the transistor 116 may serve as a pre-charge circuit topre-charge the transistors 120 and 122, more specifically, the sourcesof the transistors 120 and 122, before the transistors 120 and 122 areturned off since, at time t0, the transistors 120 and 122 are turned on.The transistor 118 may serve as a current source after the controlsignal 114's is pulled to LOW, and the transistors 124 and 130 areconfigured to perform a reset function after the control signal 114's ispulled back to HIGH, which will be discussed further below,respectively. Moreover, in some embodiments, a respective size of thetransistor 116 may be selected to be substantially smaller than othertransistors (e.g., the transistors 120, 122, 124, 126, 128, and 130)such that prior to time t0 (e.g., before signal 114′ transitions to LOW)a stand-by current (also known as a “DC current”) may be minimized andrespective logic states at nodes X and Y may remain at LOW. Thus, noiseand/or false logic state(s), caused by the latch circuit formed by thetransistors 126 and 128, can be advantageously avoided.

Subsequently, at time “t,” since both the clock signal 101 and thereference clock signal 103 have transitioned to HIGH, respectively, the(NAND) logic gate 114 transitions the control signal 114′ to LOW, whichturns off the transistor 116 and turn on the transistor 118 such thatthe transistor 116 may stop pre-charging the transistors 120 and 120 andthe transistor 118 may start charging the voltage levels at nodes X andY through the ON transistors 120 and 122, respectively. It is noted thatbecause of signal propagation delays caused by the logic gate 114, thecontrol signal 114′ may not transition to LOW immediately after bothsignals 101 and 103 transition to HIGH. As mentioned above, the firstand second delay circuits 110 and 112 respectively delay the clocksignal 101 and the reference clock signal 103. More specifically, insome embodiments, the first delay circuit 110 may delay the clock signal101 by a delay “ΔT₁” so as to provide the delayed signal 101′ as shown;and the second delay circuit 112 may delay the clock signal 103 by adelay “ΔT₂” so as to provide the delayed signal 103′ as shown. In someembodiments, the delays ΔT₁ and ΔT₂ may be substantially similar to eachother.

At time “t2,” because of the delays, rising edges of the delayed signals101′ and 103′ have not been received by the “PMOS” transistors 120 and122, i.e., the delayed signals 101′ and 103′ are still at LOW. Thus, thetransistors 120 and 122 remain in the ON state. And the transistor 116remains OFF and the transistor 118 remains ON because the control signal114′ has been pulled to LOW at time t. The transistor 118, which servesas the current source as mentioned above, is configured to keep chargingvoltage levels at nodes X and Y. As such, the voltage levels at nodes Xand Y (i.e., the signals 131 and 133) may be charged to HIGH through theON transistors 120 and 122.

At time “t3,” the rising edge of the delayed signal 101′ is received bythe gate of the transistor 120 so that the transistor 120 is turned off.Accordingly, the voltage level at the node X (i.e., the signal 131)starts being discharged through the transistor 126 at time t3.Similarly, at time “t4,” the rising edge of the delayed signal 103′ isreceived by the gate of the transistor 122 so that the transistor 122 isturned off. Accordingly, the voltage level at the node Y (i.e., thesignal 133) starts being discharged through the transistor 128 at timet4. In some embodiments, because of the substantially similar delays ΔT₁and ΔT₂, the timing difference “ΔT” between the rising edges 101 r and103 r is reflected to the delayed signals 101′ and 103′ accordingly toturn off the transistors 120 and 122 at different times. The signals 131and 133 may start being discharged at different times, i.e., the timest3 and t4 are different and the time t4 is subsequent to the time t3. Assuch, the signal 131 may transition to LOW faster than the signal 133.Moreover, as mentioned above, the transistors 126 and 128 function as alatch circuit. That is, once either one of the signals 131 and 133transitions to a detectable logic state (e.g., a low enough voltagelevel), the logic states of the signals 131 and 133 may be latched totheir current respective states. In a non-limiting example, when eitherone of the signals 131 and 133 transitions to a low enough voltagelevel, the logic state of the signal that transitions to the low enoughvoltage level may be latched to LOW, and the logic state of the othersignal may be complementarily latched to HIGH (i.e., stops beingdischarged).

In the example of FIG. 3, since the signal 131 transitions to LOW (i.e.,a low enough voltage level) at about time “t” while the signal 133 isstill being discharged, the logic states of the signals 131 and 133 maybe latched to LOW and HIGH, respectively. That is, the signal 131 islatched to LOW and the signal 133 stops being discharged and latched toHIGH. As mentioned above, in some embodiments, the tuning circuit 138determines the pre-defined timing offset window T_(os), and the timingoffset window T_(os) is used to determine whether the signals 131 and133 can be latched by the coupled latch circuit formed by thetransistors 126 and 128, as explained below.

In an example, in a scenario where the signals 131 and 133 startdischarging at the same time (i.e., t3=t4) or at two substantially closetimes (i.e., t4 is substantially close to t3), the logic states of thesignals 131 and 133 become non-differentiable (i.e., both logic statesof the signals 131 and 133 are at either HIGH or LOW), which causes thelatch circuit formed by the transistors 126 and 128 to fail to latch alogic state within such a narrow timing difference between times t3 andt4. Alternatively stated, when the timing difference between times t3and t4 becomes smaller than the timing offset window T_(os), the latchcircuit formed by the transistors 126 and 128 cannot latch signal 131and signal 133 into inversed logic states (either HIGH or LOW).

On the other hand, as shown in FIG. 3, when the timing differencebetween times t3 and t4 exceeds the timing offset window T_(os), thelogic states of the signals 131 and 133 are differentiable because thelogic state of the signal 131 transitions to LOW first. Accordingly, thelatch circuit formed by the transistors 126 and 128 can latch the logicstates of the signals 131 and 133 as LOW and HIGH, respectively.Subsequently, the signals 131 and 133 are logically inverted through therespective inverters 132 and 133 to become the signals 135 (nowtransitioning to HIGH) and 137 (now transitioning to LOW), as shown inFIG. 1.

At time “t6,” the logic gate 136 performs the XOR logic function on thelogically inverted signals 135 and 137. As known in the art, an XORlogic gate outputs a HIGH when inputs of the XOR logic gate are indifferent logic states. Accordingly, the (XOR) logic gate 136transitions the signal 105 to HIGH at time t6. As mentioned above, whenthe signal 105 is pulled to HIGH, the pJD circuit 100 may thus determinethat the deviation ΔT of the rising edge 101 r (of the clock signal 101)from the rising edge 103′ (of the reference clock signal 103) exceedsthe pre-defined timing offset window T_(os), in accordance with someembodiments.

Subsequently, at time “t7,” since at least one of the clock signal 101and the reference clock signal 103 transitioned to LOW, the controlsignal 114′ (NAND'ing at least one LOW from either the signal 101 orsignal 103) transitions to HIGH. Accordingly, the transistors 124 and130 are turned on. As mentioned above, the transistors 124 and 130, insome embodiments, may form a reset circuit. That is, when thetransistors 124 and 130 are turned on, such a reset circuit is enabled,which starts to discharge the signals 131 and 133. In some embodiments,the signal 133 may be pulled back to LOW slightly after time t7.

At time “t8,” the signals 135 and 137 both transition to HIGH bylogically inverting the signals 131 and 133 through the inverters 132and 134, respectively, so that the signal 105 is reset to LOW (XOR'ingtwo HIGH's of the signals 135 and 137). It is noted that because of somesignal propagation delays caused by the inverters 132 and 134,respectively, the signal 105 may not transition to LOW immediately afterthe signals 131 and 133 are pulled back to LOW. In some embodiments,after the signal 105 is reset to LOW, following the operations describedabove, the pJD circuit 100 may be configured to be ready to monitorwhether a subsequent rising edge (e.g., 101 r′) of the clock signal 101contains an intolerable amount of jitter when comparing to a rising edge(e.g., 103 r′) of the reference clock signal 103. The rising edge 11 r′may be received by the first delay circuit 110 at a subsequent time(e.g., time “t9”), and the rising edge 103 r′ may be received by thesecond delay circuit 112 at another subsequent time (e.g., time “t10”).

FIG. 4 illustrates an exemplary flow chart of a method 400 to operatethe pJD circuit 100 of FIG. 1, in accordance with some embodiments. Invarious embodiments, the operations of the method 400 are performed bythe respective components illustrated in FIGS. 1-3. For purposes ofdiscussion, the following embodiment of the method 400 will be describedin conjunction with FIGS. 1-3. The illustrated embodiment of the method400 is merely an example. Therefore, it should be understood that any ofa variety of operations may be omitted, re-sequenced, and/or added whileremaining within the scope of the present disclosure.

The method 400 starts with operation 402 in which a clock signal and areference clock signal are received, in accordance with variousembodiments. In the example illustrated in FIGS. 1-3, the clock signal101 and the reference clock signal 103 that present the timingdifference ΔT between their respective rising edges (101 r and 103 r)are received by the pJD circuit 100. The pJD circuit 100 is configuredto detect whether the timing difference ΔT exceeds the pre-definedtiming offset window T_(os).

The method 400 continues to operation 404 in which respective risingedges of the clock signal and reference clock signal are delayed, inaccordance with various embodiments. Continuing with the above example,the clock signal 101 is delayed by the first delay circuit 110 as thedelayed signal 101′, wherein the delayed signal 101′ is ΔT₁ behind theclock signal 101. The reference clock signal 103 is delayed by thesecond delay circuit 112 as the delayed signal 103′, wherein the delayedsignal 103′ is ΔT₂ behind the reference clock signal 103. As such, therespective rising edges 101 r and 103 r are delayed by ΔT₁ and ΔT₂,respectively. More specifically, in some embodiments, ΔT₁ and ΔT₂ aresubstantially similar to each other so that the timing difference ΔT maybe reflected to the rising edges of the delayed signal 101′ and 103′.

The method 400 continues to operation 406 in which the delayed risingedges are received by cross-coupled first and second transistors tocause the first and second transistors to be turned off, respectively,such that voltage levels at respective drains of the first and secondtransistors start being discharged, in accordance with variousembodiments. Continuing with the above example, since the delayed risingedges (i.e., the rising edges of the delayed signals 101′ and 103′)reflect the timing difference ΔT (between the rising edges 101 r and 103r), the first transistor (e.g., 120) receives the rising edge of thedelayed signal 101′ before the second transistor (e.g., 122) receivedthe rising edge of the delayed signal 103′. As such, the voltage levelof the drain of the first transistor 120 may start being dischargedbefore the voltage level of the drain of the second transistor 122starts being discharged.

The method 400 continues to operation 408 in which corresponding logicstates of the voltage levels at the drains of the first and secondtransistors are latched by the first and second transistors when thetiming difference between the rising edges of the clock signal and thereference clock signal exceeds the pre-defined timing offset windowT_(os), in accordance with various embodiments. Still continuing withthe above example, because of the timing difference ΔT, the voltagelevel of the drain of the first transistor 120 starts being dischargedfirst. The voltage level of the drain of the first transistor 120 may bedischarged low enough to reach a corresponding LOW first while thevoltage level of the drain of the second transistor 122 may be stillbeing discharged (and not low enough to reach a corresponding LOW). Assuch, the logic state at the drain of the first transistor 120 may belatched to LOW, and the logic state at the drain of the secondtransistor 122 may be complementarily latched to HIGH. In someembodiments, the timing offset window T_(os) may be pre-defined based onwhether the cross-coupled first and second transistors 120 and 122 areable to latch a logic state at one drain of the transistors 120 and 122within the timing difference ΔT. In this case, the pJD circuit 100 maydetermine that the timing difference ΔT exceeds the pre-defined timingoffset window T_(os) since the logic state at the drain of the firsttransistor 120 is able to be latched.

FIG. 5 illustrates an exemplary circuit diagram of an n-type jitterdetection circuit (hereinafter “nJD circuit”) 500, in accordance withsome embodiments. Similar to the pJD circuit 100, the nJD circuit 500 isconfigured to compare respective transition edges between a clock signal(e.g., 501) and a reference clock signal (e.g., 503) so as to determinewhether the clock signal 501 contains jitter that exceeds a pre-definedthreshold (e.g., the pre-defined timing offset window T_(os) discussedabove with respect to FIG. 3). If so, the nJD circuit 500 may outputsignal 505 at a high logic state (hereinafter “HIGH”). On the otherhand, if no jitter is detected or the jitter in the clock signal 501does not exceed the pre-defined threshold, the nJD circuit 500 mayoutput the signal 505 at a low logic state (hereinafter “LOW”).

In some embodiments, the clock signal 501 and the reference clock signal503 are substantially similar to the clock signal 101 and the referenceclock signal 103. For purposes of clarity, the clock signal and thereference clock signal will be referred to as the clock signal 501 andthe reference clock signal 503, respectively, in the followingdiscussions. Also, in some embodiments, the nJD circuit 500 issubstantially similar to the pJD circuit 100 of FIG. 1 except that thenJD circuit 500 is configured to compare respective “falling” edges ofthe clock signal 501 and the reference clock signal 503 by usingdifferent types of transistors. Thus, the nJD circuit 500 will bebriefly discussed below.

Similar to the pJD circuit 100, in some embodiments, the nJD circuit 500includes a first delay circuit 510, a second delay circuit 512, a logicgate 514, transistors 516, 518, 520, 522, 524, 526, 528, and 530,inverters 532 and 534, a logic gate 536, and a tuning circuit 538. Thetuning circuit 538 is substantially similar to the tuning circuit 138,which is described above with respect to FIG. 2. Also, the first andsecond delay circuits 510 and 512 may each include a plurality ofserially coupled buffers, inverter, or the like (not shown). The firstdelay circuit 510 is configured to receive the clock signal 501 andprovide a delayed version of the clock signal, e.g., 501′, and thesecond delay circuit 512 is configured to receive the reference clocksignal 503 and provide a delayed version of the clock signal, e.g.,503′.

Different from the pJD circuit 100, in some embodiments, the logic gate514 of the nJD circuit 500 may include a NOR logic gate that isconfigured to perform a NOR logic function on the clock signal 501 andthe reference clock signal 503 so as to provide a control signal 514′based on a NOR'ed result of logic states of the clock signal 501 and thereference clock signal 503. Further, the transistors 518, 520, and 522may be each implemented by an NMOS FET, and the transistors 516, 524,526, 528, and 530 may be each implemented by a PMOS FET. However, it isnoted that the transistors 516 to 530 may be each implemented by any ofvarious types of transistors (e.g., a bipolar junction transistor (BJT),a high-electron mobility transistor (HEMT), etc.) while remaining withinthe scope of the present disclosure.

In some embodiments, the transistors 516 and 518 are commonly coupled toa first supply voltage 507 (e.g., Vss or ground) at a respective drainand source, and gated by the control signal 514′. The transistor 520 iscoupled to the transistor 516's source by its respective source, andgated by the delayed clock signal 501′. The transistor 522 is coupled tothe transistor 518's drain by its respective source, and gated by thedelayed reference clock signal 503′. And the transistor 518's drain iscoupled to the transistor 516's source. The transistors 524 and 526 arecoupled to a drain of the transistor 520 by their respective drains at acommon node “A,” and to a second supply voltage 509 (e.g., Vdd) by theirrespective sources. In some embodiments, the transistor 524 is gated bythe control signal 514′. Similarly, the transistors 528 and 530 arecoupled to a drain of the transistor 522 by their respective drains at acommon node “B,” and to the second supply voltage 509 (e.g., Vdd) bytheir respective sources. In some embodiments, the transistor 530 isgated by the control signal 514′.

More specifically, in some embodiments, the transistors 526 and 528 arecross-coupled to each other. That is, a gate of the transistor 526 iscoupled to the drain of the transistor 528 and a gate of the transistor528 is coupled to the drain of the transistor 526 so as to allow thetransistors 526 and 528 to function as a latch circuit that issubstantially similar to the latch circuit formed by the transistors 126and 128 of the pJD circuit 100.

In some embodiments, the inverters 532 and 534 are configured to receivesignals present at nodes A and B (hereinafter “signal 531” and “signal533”), respectively, as respective input signals, and provide respectivelogically inverted signals 535 and 537. The signals 535 and 537 arereceived by the logic gate 536, which may be similarly implemented as anXOR logic gate in some embodiments. The logic gate 536 is configured toperform the XOR logic function on the signals 535 and 537 so as toprovide the signal 505 whose logic state is determined based on anXOR'ed result of logic states of the signals 535 and 537.

FIG. 6 illustrates exemplary waveforms of signals 501, 503, 514′, 501′,503′, 531, 533, and 505 to operate the nJD circuit 500 of FIG. 5, inaccordance with some embodiments. Each waveform of the signals 501, 503,514′, 501′, 503′, 531, 533, and 505 illustrated in FIG. 6 varies betweenHIGH and LOW over time.

Similar to the operation of the pJD circuit 100, in some embodiments,the reference clock signal 503 may be used as the “presumably periodicsignal,” and the clock signal 501 may be used as a to-be examined signalto determine whether a deviation of the clock signal 501 from thepresumably periodic signal 503 exceeds the pre-defined timing offsetwindow T_(os). When the clock signal 501 contains jitter (i.e., thedeviation) that exceeds the pre-defined timing offset window (i.e., anintolerable amount of jitter) on its respective falling edge, the nJDcircuit 500 may pull the signal 505 to HIGH. Accordingly, in order toexplain how the “intolerable” jitter on the falling edge of the clocksignal 501 is detected by the nJD circuit 500, in FIG. 6, the waveformof signal 501 (received by the nJD circuit 500) illustrates such ascenario and how the nJD circuit 500 responds by using signals 514′,501′, 503′, 531, and 533 to pull the signal 505 to HIGH.

As shown in FIG. 6, the clock signal 501's falling edge 501 f isdeviated from the reference clock signal 503's falling edge 503 f. Morespecifically, the falling edge 501 f occurs “ΔT” ahead of the fallingedge 503 f. Alternative stated, the rising edges 501 f and 503 f have atiming difference ΔT from each other. As described above, the logic gate514 performs the NOR logic function on the clock signal 501 and thereference clock signal 503. As known in the art, only when both thesignals 501 and 503 transition to LOW, the logic gate 514 can output thecontrol signal 514′ as HIGH.

Prior to time “t0,” the control signal 514′ is at LOW, and at time t0,the control signal 514′ remains at LOW, because the logic states of thesignals 501 and 503 are at HIGH. It is noted that the transistors 516,518, 524, and 530 are all gated by the signal 514′. Accordingly, whenthe control signal 514′ is at LOW, the “NMOS” transistor 518 is turnedoff, and the “PMOS” transistors 516, 524, and 530 are turned on. In someembodiments, the transistor 516 may serve as a pre-discharge circuit topre-discharge the transistors 520 and 522, more specifically, thesources of the transistors 520 and 522, before the transistors 520 and522 are turned off, since, at time t0, the transistors 520 and 522 areturned on. The transistor 518 may serve as a current sink after thecontrol signal 514's is pulled to HIGH, and the transistors 524 and 530are configured to perform a reset function after the control signal514's is pulled back to LOW, which will be discussed below,respectively. Moreover, in some embodiments, a respective size of thetransistor 516 may be selected to be substantially smaller than othertransistors (e.g., the transistors 520, 522, 524, 526, 528, and 530)such that prior to time t0 (e.g., before signal 514′ transitions toHIGH) a stand-by current (also known as a “DC current”) may be minimizedand respective logic states at nodes A and B may remain at HIGH. Thus,noise and/or false logic state(s), caused by the latch circuit formed bythe transistors 526 and 528, can be advantageously avoided.

Subsequently, at time “t1,” since both the clock signal 501 and thereference clock signal 503 have transitioned to LOW, respectively, the(NOR) logic gate 514 transitions the control signal 514′ to HIGH, whichturns off the transistor 516 and turn on the transistor 518 such thatthe transistor 516 may stop pre-discharging the transistors 520 and 520and the transistor 518 may start discharging the voltage levels at nodesA and B through the ON transistors 520 and 522, respectively. It isnoted that because of a signal propagation delay caused by the logicgate 514, the control signal 514′ may not transition to HIGH immediatelyafter both signals 501 and 503 transition to LOW. As mentioned above,the first and second delay circuits 510 and 512 respectively delay theclock signal 501 and the reference clock signal 503. More specifically,in some embodiments, the first delay circuit 510 may delay the clocksignal 501 by a delay “ΔT” so as to provide the delayed signal 501′ asshown; and the second delay circuit 512 may delay the clock signal 503by a delay “ΔT₂” so as to provide the delayed signal 503′ as shown. Insome embodiments, the delays ΔT₁ and ΔT₂ may be substantially similar toeach other.

At time “t2,” because of the delays, falling edges of the delayedsignals 501′ and 503′ have not been received by the “NMOS” transistors520 and 522, i.e., the delayed signals 501′ and 503′ are still at HIGH.Thus, the transistors 520 and 522 are remained ON. And the transistor516 is remained OFF and the transistor 518 is remained ON because thecontrol signal 514′ has been pulled to HIGH at time t1. The transistor518, served as the current sink as mentioned above, is configured tokeep discharging voltage levels at nodes A and B. As such, the voltagelevels at nodes A and B (i.e., the signals 531 and 533) may bedischarged to LOW through the ON transistors 520 and 522.

At time “t3,” the falling edge of the delayed signal 501′ is received bythe gate of the transistor 520 so that the transistor 520 is turned off.Accordingly, the voltage level at the node A (i.e., the signal 531)starts being charged through the transistor 526 at time t3. Similarly,at time “t4,” the falling edge of the delayed signal 503′ is received bythe gate of the transistor 522 so that the transistor 522 is turned off.Accordingly, the voltage level at the node B (i.e., the signal 533)starts being charged through the transistor 528 at time t4.

In some embodiments, because of the substantially similar delays ΔT₁ andΔT₂, the timing difference “ΔT” between the falling edges 501 f and 503f is reflected to the delayed signals 501′ and 503′ accordingly to turnoff the transistors 520 and 522 at different times. The signals 531 and533 may start being charged at different times, i.e., the times t3 andt4 are different and the time t4 is subsequent to the time t3. As such,the signal 531 may transition to HIGH faster than the signal 533.Moreover, as mentioned above, the transistors 526 and 528 function as alatch circuit. That is, once either one of the signals 531 and 533transitions to a detectable logic state (e.g., a high enough voltagelevel), the logic states of the signals 531 and 533 may be latched aswhat they currently are. In a non-limiting example, when either one ofthe signals 531 and 533 transitions to a high enough voltage level, thelogic state of the signal that transitions to the high enough voltagelevel may be latched to HIGH, and the logic state of the other signalmay be complementarily latched to LOW (i.e., stops being charged).

In the example of FIG. 5, since the signal 531 transitions to HIGH(i.e., a high enough voltage level) at about time “t5” while the signal533 is still being charged, the logic states of the signals 531 and 533may be latched to HIGH and LOW, respectively. That is, the signal 531 islatched to HIGH and the signal 533 is stopped being charged and latchedto LOW. And as mentioned above, in some embodiments, the tuning circuit538 determines the pre-defined timing offset window T_(os), and thetiming offset window T_(os) is used to determine whether the signals 531and 533 can be latched by the coupled latch circuit formed by thetransistors 526 and 528.

In an example, when the signals 531 and 533 start being charged at thesame time (i.e., t3=t4) or at two substantially close times (i.e., t4 issubstantially close to t3), the logic states of the signals 531 and 533become non-differentiable (i.e., both logic states of the signals 531and 533 are at either HIGH or LOW), which causes the latch circuitformed by the transistors 526 and 528 to fail to latch a logic statewithin such a narrow timing difference between times t3 and t4.Alternatively stated, when the timing difference between times t3 and t4becomes smaller than the timing offset window T_(os), the latch circuitformed by the transistors 526 and 528 cannot latch signal 531 and signal533 into inversed logic states (either HIGH or LOW).

On the other hand, which is the case shown in FIG. 6, when the timingdifference between times t3 and t4 exceeds the timing offset windowT_(os), the logic states of the signals 531 and 533 are differentiablebecause the logic state of the signal 531 transitions to HIGH first.Accordingly, the latch circuit formed by the transistors 526 and 528 canlatch the logic states of the signals 531 and 533 as HIGH and LOW,respectively. Subsequently, the signals 531 and 533 are logicallyinverted through the respective inverters 532 and 534 to become thesignals 535 (now transitioning to LOW) and 537 (now transitioning toHIGH).

At time “t6,” the logic gate 536 performs the XOR logic function on thelogically inverted signals 535 and 537. As described above, an XOR logicgate outputs a HIGH when inputs of the XOR logic gate are in differentlogic states. Accordingly, the (XOR) logic gate 536 transitions thesignal 505 to HIGH at time t6. When the signal 505 is pulled to HIGH,the nJD circuit 500 may thus determine that the deviation ΔT of therising edge 501 f (of the clock signal 501) from the rising edge 503′(of the reference clock signal 503) exceeds the pre-defined timingoffset window T_(os), in accordance with some embodiments.

Subsequently, at time “t7,” since at least one of the clock signal 501and the reference clock signal 503 transitioned to HIGH, the controlsignal 514′ (NOR'ing at least one HIGH from either the signals 501 orsignal 503) transitions to LOW. Accordingly, the transistors 524 and 530are turned on. As mentioned above, the transistors 524 and 530, in someembodiments, may form a reset circuit. That is, when the transistors 524and 530 are turned on, such a rest circuit is enabled, which starts tocharge the signals 531 and 533. In some embodiments, the signal 533 maybe pulled back to HIGH slightly after time t7.

At time “t8,” the signals 535 and 537 both transition to LOW bylogically inverting the signals 531 and 533 through the inverters 532and 534, respectively, so that the signal 505 is reset to LOW (XOR'ingtwo LOW's of the signals 535 and 537). It is noted that because of somesignal propagation delays caused by the inverters 532 and 534,respectively, the signal 505 may not transition to LOW immediately afterthe signals 531 and 533 are pulled back to HIGH. In some embodiments,after the signal 505 is reset to LOW, following the operations describedabove, the nJD circuit 500 may be configured to be ready to monitorwhether a subsequent falling edge (e.g., 501 f′) of the clock signal 501contains an intolerable amount of jitter when comparing to a fallingedge (e.g., 503 f′) of the reference clock signal 503. The falling edge501 f may be received by the first delay circuit 510 at a subsequenttime (e.g., time “t9”), and the falling edge 503 f may be received bythe second delay circuit 512 at another subsequent time (e.g., time“t10”).

FIG. 7 illustrates an exemplary flow chart of a method 700 to operatethe nJD circuit 500 of FIG. 5, in accordance with some embodiments. Invarious embodiments, the operations of the method 700 are performed bythe respective components illustrated in FIGS. 2, and 5-6. For purposesof discussion, the following embodiment of the method 700 will bedescribed in conjunction with FIGS. 2, and 5-6. The illustratedembodiment of the method 700 is merely an example. Therefore, it shouldbe understood that any of a variety of operations may be omitted,re-sequenced, and/or added while remaining within the scope of thepresent disclosure.

The method 700 starts with operation 702 in which a clock signal and areference clock signal are received, in accordance with variousembodiments. In the example illustrated in FIGS. 5-6, the clock signal501 and the reference clock signal 503 that present the timingdifference ΔT between their respective falling edges (501 f and 503 f)are received by the nJD circuit 500. The nJD circuit 500 is configuredto detect whether the timing difference ΔT exceeds the pre-definedtiming offset window T_(os).

The method 700 continues to operation 704 in which respective fallingedges of the clock signal and reference clock signal are delayed, inaccordance with various embodiments. Continuing with the above example,the clock signal 501 is delayed by the first delay circuit 510 as thedelayed signal 501′, wherein the delayed signal 501′ is ΔT₁ behind theclock signal 501. The reference clock signal 503 is delayed by thesecond delay circuit 512 as the delayed signal 503′, wherein the delayedsignal 503′ is ΔT₂ behind the reference clock signal 503. As such, therespective falling edges 501 f and 503 f are delayed by ΔT₁ and ΔT₂,respectively. More specifically, in some embodiments, ΔT₁ and ΔT₂ aresubstantially similar to each other so that the timing difference ΔT maybe reflected to the falling edges of the delayed signal 501′ and 503′.

The method 700 continues to operation 706 in which the delayed fallingedges are received by cross-coupled first and second transistors tocause the first and second transistors to be turned off, respectively,such that voltage levels at respective drains of the first and secondtransistors start being charged, in accordance with various embodiments.Continuing with the above example, since the delayed falling edges(i.e., the falling edges of the delayed signals 501′ and 503′) reflectthe timing difference ΔT (between the falling edges 501 r and 503 r),the first transistor (e.g., 520) receives the falling edge of thedelayed signal 501′ before the second transistor (e.g., 522) receivedthe falling edge of the delayed signal 503′. As such, the voltage levelof the drain of the first transistor 520 may start being charged beforethe voltage level of the drain of the second transistor 522 starts beingdischarged.

The method 700 continues to operation 708 in which corresponding logicstates of the voltage levels at the drains of the first and secondtransistors are latched by the first and second transistors when thetiming difference between the falling edges of the clock signal and thereference clock signal exceeds the pre-defined timing offset windowT_(os), in accordance with various embodiments. Still continuing withthe above example, because of the timing difference ΔT, the voltagelevel of the drain of the first transistor 520 starts being chargedfirst. The voltage level of the drain of the first transistor 520 may becharged high enough to reach a corresponding HIGH first while thevoltage level of the drain of the second transistor 522 may be stillbeing charged (and not high enough to reach a corresponding HIGH). Assuch, the logic state at the drain of the first transistor 520 may belatched to HIGH, and the logic state at the drain of the secondtransistor 522 may be complementarily latched to LOW. In someembodiments, the timing offset window T_(os) may be pre-defined based onwhether the cross-coupled first and second transistors 520 and 522 areable to latch a logic state at one drain of the transistors 520 and 522within the timing difference ΔT. In this case, the nJD circuit 500 maydetermine that the timing difference ΔT exceeds the pre-defined timingoffset window T_(os) since the logic state at the drain of the firsttransistor 520 is able to be latched.

In an embodiment, a noise detection circuit includes a first transistorconfigured to receive a delayed version of a clock signal; a secondtransistor configured to receive a delayed version of a reference clocksignal; and a latch circuit, coupled to the first transistor at a firstnode and coupled to the second transistor at a second node, andconfigured to latch logic states of voltage levels at the first andsecond nodes, respectively, based on whether a timing difference betweentransition edges of the clock signal and the reference clock signalexceeds a pre-defined timing offset threshold.

In another embodiment, a noise detection circuit includes a firsttransistor configured to receive a delayed version of a clock signal; asecond transistor configured to receive a delayed version of a referenceclock signal; a latch circuit, coupled to the first transistor at afirst node and coupled to the second transistor at a second node, andconfigured to latch logic states of voltage levels at the first andsecond nodes, respectively, based on whether a timing difference betweentransition edges of the clock signal and the reference clock signalexceeds a pre-defined timing offset threshold; and a plurality ofcapacitors coupled between the first and second nodes.

Yet in another embodiment, a method includes receiving a clock signaland a reference clock signal, wherein at least a transition edge of theclock signal is deviated from a transition edge of the reference clocksignal by a timing difference; delaying the clock signal and thereference clock signal; receiving the delayed clock signal and referenceclock signal by a first transistor and a second transistor,respectively, so as to start either discharging or charging voltagelevels at drains of the first and second transistors at different times;and latching respective logic states of the voltage levels at drains ofthe first and second transistors when the timing difference is greaterthan a pre-defined timing offset threshold.

The foregoing outlines features of several embodiments so that thoseordinary skilled in the art may better understand the aspects of thepresent disclosure. Those skilled in the art should appreciate that theymay readily use the present disclosure as a basis for designing ormodifying other processes and structures for carrying out the samepurposes and/or achieving the same advantages of the embodimentsintroduced herein. Those skilled in the art should also realize thatsuch equivalent constructions do not depart from the spirit and scope ofthe present disclosure, and that they may make various changes,substitutions, and alterations herein without departing from the spiritand scope of the present disclosure.

What is claimed is:
 1. A noise detection circuit, comprising: a firstcomponent configured to receive a clock signal; a second componentconfigured to receive a reference clock signal; and a latch circuit,coupled to the first component at a first node and coupled to the secondcomponent at a second node, and configured to latch logic states ofvoltage levels at the first and second nodes, respectively, based on atiming difference between transition edges of the clock signal and thereference clock signal.
 2. The circuit of claim 1, wherein the first andsecond components each comprises a p-type metal-oxide-semiconductor(PMOS) field-effect-transistor (FET).
 3. The circuit of claim 1, whereinthe latch circuit comprises a first n-type metal-oxide-semiconductor(NMOS) field-effect-transistor (FET) and a second NMOS FET that arecross-coupled to each other.
 4. The circuit of claim 3, wherein thefirst node is coupled to a source of the first transistor, a drain ofthe first NMOS FET, and a gate of the second NMOS FET, and the secondnode is coupled to a source of the second transistor, a drain of thesecond NMOS FET, and a gate of the first NMOS FET.
 5. The circuit ofclaim 1, wherein when a timing difference between rising edges of theclock signal and the reference clock signal exceeds a pre-defined timingoffset threshold, the latch circuit is configured to latch the logicstates of the voltage levels at the first and second nodes.
 6. Thecircuit of claim 5, wherein the pre-defined timing offset threshold istunable based on a number of capacitors coupled between the first andsecond nodes.
 7. The circuit of claim 5, wherein the logic states of thevoltage levels at the first and second nodes are complementary to eachother.
 8. The circuit of claim 1, further comprising: a first delaycircuit configured to provide a delayed version of the clock signal tothe first component; and a second delay circuit configured to provide adelayed version of the reference clock signal to the second component.9. The circuit of claim 1, further comprising: a NAND logic gateconfigured to perform a NAND logic function on the clock signal and thereference clock signal so as to provide a control signal.
 10. Thecircuit of claim 9, further comprising: a pre-charge transistorconfigured to pre-charge the first and second transistors when thecontrol signal is at a high logic state, and when the first and secondtransistors are turned on; and a current-source transistor configured tocharge the voltage levels at the first and second nodes when the controlsignal is at a low logic state, and when the first and secondtransistors are turned on.
 11. The circuit of claim 10, wherein thepre-charge transistor includes an n-type metal-oxide-semiconductor(NMOS) field-effect-transistor (FET), and the current-source transistorincludes a p-type metal-oxide-semiconductor (PMOS) FET.
 12. A noisedetection circuit, comprising: a first component configured to receive aclock signal; a second component configured to receive a reference clocksignal; a latch circuit, coupled to the first component at a first nodeand coupled to the second component at a second node, and configured tolatch logic states of voltage levels at the first and second nodes,respectively, based on a timing difference between transition edges ofthe clock signal and the reference clock signal; and a plurality ofcapacitors coupled between the first and second nodes.
 13. The circuitof claim 1, wherein the first and second components each comprises ann-type metal-oxide-semiconductor (NMOS) field-effect-transistor (FET).14. The circuit of claim 13, wherein the latch circuit comprises a firstp-type metal-oxide-semiconductor (PMOS) field-effect-transistor (FET)and a second PMOS FET that are cross-coupled to each other.
 15. Thecircuit of claim 14, wherein the first node is commonly coupled by adrain of the first transistor, a drain of the first PMOS FET, and a gateof the second PMOS FET, and the second node is commonly coupled by adrain of the second transistor, a drain of the second PMOS FET, and agate of the first PMOS FET.
 16. A method, comprising: receiving a clocksignal and a reference clock signal, wherein at least a transition edgeof the clock signal is deviated from a transition edge of the referenceclock signal by a timing difference; receiving the clock signal andreference clock signal by a first transistor and a second transistor,respectively, so as to start either discharging or charging voltagelevels at drains of the first and second transistors at different times;and latching respective logic states of the voltage levels at drains ofthe first and second transistors when the timing difference is greaterthan a pre-defined timing offset threshold.
 17. The method of claim 16,wherein the logic states of the voltage levels are complementary to eachother.
 18. The method of claim 16, further comprising: delaying clocksignal; and delaying the reference clock signal.
 19. The method of claim16, further comprising: performing a NOR logic function on the clocksignal and the reference clock signal so as to provide a control signal.20. The method of claim 19, further comprising: pre-discharging thefirst and second transistors when the control signal is at a low logicstate, and when the first and second transistors are turned on; anddischarging the voltage levels at the first and second nodes when thecontrol signal is at a high logic state, and when the first and secondtransistors are turned on.